The present invention relates to a frequency-shift-keying demodulator comprising the features of the preamble of independent claim 1 and to a method of frequency-shift-keying demodulating an input signal comprising the steps of the preamble of independent claim 7.
In order to reduce the power consumption of a power amplifier for wireless communication from a transmitter side and to realise bandwidth efficient wireless transmission, various continuous phase frequency-shift-keying (FSK) type modulation schemes are known. For example, minimum shift keying (MSK) and Gaussian frequency shift keying (GFSK) are widely used in wireless communication systems, e.g. according to the GSM or Bluetooth standard.
With respect to demodulation of a signal, frequency discriminators or FM-AM converters are the most popular forms of demodulators due to their simplicity.
In the following, a receiver and a demodulator for frequency-shift-keying modulation according to the prior art is described by reference to FIGS. 1 and 2.
FIG. 1 shows a conventional receiver architecture comprising an antenna 20 for reception of transmitted signals.
The antenna 20 is connected via an RF-bandpass filter 16 and a low-noise-amplifier 17 to a mixer 21.
The mixer 21 multiplies the signal received by the antenna 20 with a signal generated by a local oscillator LO to downconvert the signal to intermediate frequency (IF).
The IF signal further goes from the mixer 21 via a channel selection filter 18 and a further amplifier 19 to a conventional frequency-shift-keying FSK demodulator 11. The frequency-shift-keying demodulator 11 receives a FSK input signal i(t).
The differential frequency-shift-keying demodulator 11 demodulates the FSK input signal i(t) and outputs a low-pass filtered output signal rLP(t).
FIG. 2 shows the internal structure of a differential frequency-shift-keying demodulator according to the prior art.
The conventional differential frequency-shift-keying demodulator 11 shown in FIG. 2 consists of a phase shifter 12, a mixer 13 and a low-pass filter 15.
The input signal i(t) is supplied to both the phase shifter 12 and the mixer 13.
The phase shifter 12 shifts the phase of the input signal i(t) by a predetermined degree and outputs a shifted signal Id(t) to the mixer 13.
The mixer 13 multiplies the input signal i(t) and the shifted signal Id(t) and outputs a mixed signal r(t) to the low-pass filter 15.
The low-pass filter 15 low-pass filters the mixed signal r(t) and outputs a low-pass filtered signal rLP(t).
The FSK input signal i(t) of the above described differential frequency-shift-keying demodulator 11 can be written as
            i      ⁡              (        t        )              =                            s          ⁡                      (            t            )                          +                  n          ⁡                      (            t            )                              =                        A          ⁢                                          ⁢                      cos            ⁡                          [                                                2                  ⁢                  π                  ⁢                                                                          ⁢                                      f                    i                                    ⁢                  t                                +                                  2                  ⁢                  π                  ⁢                                                                          ⁢                  h                  ⁢                                                            ∫                                              -                        ∞                                            t                                        ⁢                                                                  m                        ⁡                                                  (                          τ                          )                                                                    ⁢                                                                                          ⁢                                              ⅆ                        τ                                                                                                        ]                                      +                  n          ⁡                      (            t            )                                ,wherein A is the amplitude, fi is the carrier frequency, m(τ) is the filtered data signal, n(t) is additive white Gaussian noise (AWGN) and h is the modulation index of the signal i(t).
The modulation index h of the signal i(t) is defined as h=2fdTs where fd is the frequency deviation and Ts is the symbol duration.
The goal of the above-described differential frequency-shift-keying FSK demodulator 11 is to provide an output signal rLP(t) that is proportional to the instantaneous frequency of the input signal i(t).
The frequency response of the phase shifter 12 of the FSK demodulator 11 isφ(f)=−π/2+2πK(f−fi).
Thus, the output Id(t) of the phase shifter 12 is
      Id    ⁡          (      t      )        =                              s          d                ⁡                  (          t          )                    +                        n          2                ⁡                  (          t          )                      =                  A        ⁢                                  ⁢                  cos          ⁡                      [                                          2                ⁢                π                ⁢                                                                  ⁢                                  f                  i                                ⁢                t                            +                              2                ⁢                π                ⁢                                                                  ⁢                h                ⁢                                                      ∫                                          -                      ∞                                        t                                    ⁢                                                            m                      ⁡                                              (                        τ                        )                                                              ⁢                                                                                  ⁢                                          ⅆ                      τ                                                                                  -                              π                /                2                            +                              2                ⁢                π                ⁢                                                                  ⁢                                  Khm                  ⁡                                      (                    t                    )                                                              +                                                n                  1                                ⁡                                  (                  t                  )                                                      ]                              +                        n          2                ⁡                  (          t          )                    where n1(t) and n2(t) are the noise terms generated by the phase shifter 12 due to the co-existence of n(t) in the input signal i(t).
The low-pass filtered signal rLP(t) output by the low-pass filter 15 after eliminating the double frequency terms is
            r      LP        ⁡          (      t      )        =            (                                                  A              2                        2                    ⁢                      sin            ⁡                          [                                                2                  ⁢                  π                  ⁢                                                                          ⁢                                      Khm                    ⁡                                          (                      t                      )                                                                      +                                                      n                    1                                    ⁡                                      (                    t                    )                                                              ]                                      +                  [                                                    s                ⁡                                  (                  t                  )                                            ×                                                n                  2                                ⁡                                  (                  t                  )                                                      +                                                            s                  d                                ⁡                                  (                  t                  )                                            ×                              n                ⁡                                  (                  t                  )                                                      +                                          n                ⁡                                  (                  t                  )                                            ×                                                n                  2                                ⁡                                  (                  t                  )                                                              ]                    )        ⁢          ❘      LP        .  
When the term 2πKhm(t) is small and the noise terms are negligible, the resulting low-pass filtered signal rLP(t) is
            r      LP        ⁡          (      t      )        ≈                              A          2                2            ·      2        ⁢    π    ⁢                  ⁢                  Khm        ⁡                  (          t          )                    .      
Thus, the transmitted data comprised in the input signal i(t) can be recovered correctly.
Corresponding differential frequency-shift-keying demodulators are described in the paper “A 1.8-V Operation RF CMOS Transceiver for 2.4-GHz-Band GFSK Applications” of Hiroshi Komurasaki, Tomohiro Sano et al. The paper was published in the IEEE journal of solid-state circuits, volume 38, no. 5, May 2003.
The above paper describes a single-chip RF transceiver LSI for 2.4-GHz-band Gaussian frequency-shift-keying applications, such as Bluetooth. This chip uses a 0.18 μm bulk CMOS process for lower current consumption. The LSI consists of almost all the required RF and IF building blocks, a transmit/receive antenna switch, a power amplifier, a low-noise-amplifier, an image rejection mixer, channel-selection filters, a limiter, a received signal strength indicator, a frequency discriminator, a voltage controlled oscillator, and a phase-locked loop synthesizer. The bandpass filter for channel selection operates at a low supply voltage. However, because large interference is roughly rejected at the output of the image rejection mixer and a wide-input-range bandpass filter with an optimized input bias is realized, the transceiver can operate at a supply voltage of 1.8 V.
Another differential frequency-shift-keying demodulator according to the prior art is described in the paper “A Low-Power CMOS Bluetooth RF Transceiver With A Digital Offset Cancelling DLL-Based GFSK Demodulator” published by Sangjin Byun, Yongchul Song, et al in the IEEE journal of solid-state circuits, volume 38, no. 10, October 2003.
This paper presents a fully integrated 0.18 μm CMOS Bluetooth transceiver. The chip consumes 33 mA in receive mode and 25 mA in transmit mode from a 3-V system supply. The receiver uses a low-IF (3-MHz) architecture, and the transmitter uses a direct modulation with ROM-based Gaussian low-pass filter and I/Q direct digital frequency synthesizer for high level of integration and low power consumption. A new frequency shift keying demodulator based on a delay-locked loop with a digital frequency offset canceller is proposed. The demodulator operates without harmonic distortion, handles up to ±160-kHz frequency offset, and consumes only 2 mA from a 1.8-V supply. The receiver dynamic range is from −78 dBm to −16 dBm at 0.1% bit-error rate, and the transmitter delivers a maximum of 0 dBm with 20-dB digital power control capacity.
It is a disadvantage with the above-described differential frequency-shift-keying demodulators according to the prior art that an active mixer is required to achieve a sufficient signal strength of the FSK signal. The active mixer requires a strong local oscillator (LO) signal that consumes much power for high frequencies (e.g. 60 GHz) applications.
Furthermore, mixers working at high frequencies (e.g. 60 GHz) are very complicated and thus expensive elements.